For many years frequency division duplex (FDD) and time division duplex (TDD) have been the two methods of choice for handling uplink (UL) and downlink (DL) transmissions in wireless systems. FDD uses two different frequencies for the UL and DL thereby separating them in frequency whereas TDD utilises a single frequency for both UL and DL signals and separates them in time. Therefore, in order to meet the performance requirements of a number of telecommunications standards, integrated circuits (ICs) and/or communication units have been designed to utilise frequency division duplex (FDD) techniques as a mechanism to separate UL/DL transmit and receive communications.
As a consequence, and particularly at typical wireless communication frequencies where the transmit (and therefore receive) frequency is very high, such as in the 1 GHz frequency region in the third generation (3G) wideband code division multiple access (WCDMA) standard, it is known that interference is caused by poor isolation between the transmitted and received signals at these very high frequencies within the ICs or communication units. Here, the transmit signal leaks through the duplex filter and mixes via the mechanism of second order distortion within the receive mixer to baseband, thereby resulting in degraded receive signal to noise ratio (SNR) performance. This causes an effective desensitization of the receiver. The problem becomes critical when the transmitter is operating at, or near, the transmitter's maximum transmit power capability whilst the receiver is operating at, or near, its minimum receive power capability, referred to as the receiver's ‘sensitivity’. In this scenario, such 2nd order intermodulation products can ‘desense’ the radio and lead to bit-error-rate (BER) failure. Second order intermodulation distortion (IM2 or IIP2) occurs when two signals mix with each other through a second order nonlinearity to produce an intermodulation product at the sum and difference frequencies of the two interferers.
FIG. 1 schematically illustrates known circuitry and a cause of such 2nd order intermodulation product interference effects in a high frequency communication unit 100. The high frequency communication unit 100 comprises digital baseband ‘I’ and ‘Q’ signals 102 being input to a transmit digital-to-analogue converter (TX DAC) 105, where the digital baseband ‘I’ and ‘Q’ signals 102 are converted to analogue baseband ‘I’ and ‘Q’ signals and filtered in low pass filter (LPF) 110. The filtered baseband signals are then up-converted in frequency using a mixer stage 115 coupled to a local oscillator (LO) 120, such that the filtered baseband signals are translated in frequency to the frequency of the LO signal provided the LO 120. The up-converted signal output from the mixer stage 115 is input to a power amplifier (PA) 125, where it is amplified to a sufficiently high radio frequency level to be radiated from antenna 135. The antenna 135 is coupled to a (transmit (Tx)/receive (Rx)) duplex filter 130 which attempts to attenuate signals received from the transmit path from entering the communication unit's receive path. However, given the limitations of filtering technology at such high radio frequencies, a significant amount of the transmit signal is leaked 140 into the receiver path.
Thus, in the receive path, the antenna 135 and Tx/Rx duplex filter 130 route received high frequency signals to a low noise amplifier (LNA) 145. The amplified high frequency signal is input to a quadrature down-mixer 150, which down-converts the amplified signal by multiplying it with a quadrature shifted 155 local oscillator (LO) signal that is fed from a LO source 160. The outputs from the quadrature down-mixer 150 are at baseband frequencies, such that low-pass or band-pass filtering (LPF/BPF) 165 can be used to remove or attenuate undesired signals in the frequency domain. The baseband signals may be at a low frequency (LF) signal, a very low intermediate frequency (VLIF) signal or even a DC (zero IF) signal. Baseband (analogue) filtered signals are then digitised in the receive analogue-to-digital converter (RX ADC) 170 and filtered again to remove quantization effects in filter 175. Graph 185 illustrates how the performance of the receiver is de-sensitised (often referred to as ‘desense’) by the leakage of the transmit signal into the receive path, with most of the desense effect occurring in the receive down-mixer stage. The performance reduction is measured in terms of ‘desense’ and ultimately bit error rate (BER) and is due to a presence of IMD2 products in the baseband signal.
The classical solution to minimising the level of transmit signal leaking into the receive path uses a surface acoustic wave (SAW) filter. However, the use of SAW filters is no longer acceptable due to their large size and high cost factors, coupled to the ever-increasing need to minimise product cost and size, particularly in the mobile telephone handset business.
One attempted solution has been to use an integrated narrow bandwidth, tuneable band-pass or notch type filter, to replace the functionality of the SAW filter. However, this solution suffers from the need to use multiple lumped element inductors.
A yet further alternative approach is to employ a calibration scheme that trims the receive down-mixer operation for maximum IIP2 across process, voltage and temperature (PVT) variation. However, it is believed that this approach may not be effective, as analogue radio frequency designs do not cope well across PVT. Furthermore, there is a concern that the addition of a dedicated trimming port could potentially compromise other key RF metrics.
One known example of cancellation of second order intermodulation products is illustrated in FIG. 2. As shown, the digital baseband ‘I’ and ‘Q’ signals 102 are also input to an adaptive intermodulation distortion (IMD) cancellation function 215. The IMD cancellation function 215 is arranged to provide a digital estimate (in signals 220, 225) of the transmitter second order intermodulation distortion components of the communication unit. Thereafter, the digital estimate (in signals 220, 225) of the transmit second order intermodulation distortion components are subtracted from the signals output from filter 175 in subtraction blocks 230, 235, thereby (in principle) removing a portion of the second order intermodulation distortion components that have been created in the receive path as a result of leakage of the transmit signal through the duplex filter 130. In this manner, the estimation of interference is based on a correlated reference. Thus, the technique of FIG. 2 generates an error signal after DC correction of the receive signal, and uses this error signal to train the adaptive interference cancellation in IMD cancellation function 215. Thereafter, the cancellation is adapted so as to minimise the mean squared power of the estimation error.
There are some disadvantages that exist in the related art, such as the selectivity of the receiver performance (when a bandwidth (BW) is, say, <100 Hz) comes at a disadvantage of too slow a settling time, for example due to any averaging technique used. Hence, a designer in considering DC correction techniques is confronted with a trade off between selectivity versus settling time. A further disadvantage that exists in the related art is that a common ‘I’ and ‘Q’ path and hence a single gain stage are used to control the cancellation signal. A yet further disadvantage that exists in the related art is that a fixed (and therefore rigid) value of adaptation rate is selected to work across the power range of the communication unit.
Thus, a need exists for improved integrated circuits, communication units and methods of cancellation therefor.